Vector control device of induction motor, vector control method of induction motor, and drive control device of induction motor

ABSTRACT

The vector control device includes: secondary magnetic flux command computing means ( 40 ) for computing a secondary magnetic flux command to an induction motor ( 6 ) by taking a maximum voltage that an inverter ( 4 ) can generate into account on a basis of a torque command from an external, a DC voltage to be inputted into the inverter, and an inverter angular frequency, which is an angular frequency of an AC voltage to be outputted from the inverter; q-axis/d-axis current command generating means ( 8  and  9 ) for generating a q-axis current command and a d-axis current command on a d-q axes rotating coordinate system in reference to a secondary magnetic flux of the induction motor ( 6 ) on a basis of the torque command and the secondary magnetic flux command; output voltage computing means (voltage non-interference computation portion  14 , adder  17 , and adder  18 ) for computing an output voltage that the inverter ( 4 ) is to output on a basis of the q-axis current command, the d-axis current command, and a circuit constant of the induction motor ( 6 ); and voltage command/PWM signal generating means ( 50 ) for controlling the inverter ( 4 ) for the inverter ( 4 ) to output the output voltage.

TECHNICAL FIELD

The present invention relates to a vector control device of an inductionmotor connected to an inverter that converts a DC voltage to an ACvoltage at an arbitrary frequency to output the AC voltage, a vectorcontrol method of an induction motor, and a drive control device of aninduction motor.

BACKGROUND ART

A basic technique of vector control on an induction motor using aninverter is a prior art that has been used extensively in the industrialfield. This technique is to control a torque of a motor instantaneouslyat a high speed through an operation of a torque component current in anorthogonal relation to a secondary magnetic flux inside the motor byoperating the magnitude and the phase of an inverter output voltageseparately.

The vector control of an induction motor is a technique that is beingused also in the electric railroad in recent years.

A driving inverter of an electric vehicle is characterized in that theswitching mode of the inverter is switched in such a manner that amulti-pulse PWM mode, which is employed generally in many cases, is usedin a low speed range and a single-pulse mode is used in a medium andhigh speed range in which the inverter output voltage saturates and isfixed to the maximum value.

The multi-pulse PWM (pulse width modulation) mode referred to herein isa generally well-known PWM method and it is a mode to generate a PWMsignal by comparing a triangular wave at a frequency of about 1 kHz witha voltage command.

The single-pulse mode referred to herein is to shape an outputline-to-line voltage of the inverter to the waveform of 120°rectangular-wave conduction. Because the effective value of thefundamental wave of an inverter output voltage can be increased to themaximum and the number of pulses in a half cycle of the output voltagefundamental wave can be reduced to one, which is the minimum, it ischaracterized in that a compact and light inverter can be obtained byminimizing a switching loss of the inverter and making a cooling devicesmaller.

The waveform of 120° rectangular-wave conduction referred to herein is avoltage waveform by which a line-to-line voltage of the inverter has onepulse in a half cycle and a conduction width is 120° in electric angle.

For the inverter in an electric vehicle, it is essential to have thecapability of performing stable vector control over the entire rangefrom the multi-pulse PWM mode in a low speed range to the single-pulsemode in a medium and high speed range in which an output voltage of theinverter saturates and is fixed to the maximum value, and a vectorcontrol technique in an output voltage saturation range of the inverterand a pulse mode switching technique are crucial elements.

In particular, the magnitude of an output voltage of the inverter isfixed to the maximum voltage corresponding to an input voltage of theinverter in the output voltage saturation range of the inverter. It istherefore necessary to devise a technique to establish vector control.

In the output voltage saturation range of the inverter, in a case wherean inverter output voltage command computed by a vector control deviceexceeds the maximum voltage that the inverter can actually output, theinverter fails to output a voltage according to the inverter outputvoltage command.

Accordingly, there is a discrepancy between a secondary magnetic fluxcommand to the induction motor and a secondary magnetic flux inside themotor, which makes it difficult to perform vector control appropriately.

In order to avoid such a phenomenon, it is necessary to adjust asecondary magnetic flux command so that an inverter output voltagecommand will not exceed the maximum voltage that the inverter canactually output.

To be more concrete, in a case where the inverter output voltage commandexceeds the maximum voltage that the inverter can actually output, theinverter output voltage command has to be lowered by lowering thesecondary magnetic flux command.

Non-Patent Document 1 specified below discloses a vector control methodthat solves the problems discussed above.

Non-Patent Document 1 discloses that the inverter output voltage commandcan be corrected so as to coincide with the maximum output voltage thatthe inverter can actually output and hence vector control is enabledeven in the output voltage saturation range of the inverter byconfiguring in such a manner that when an inverter output voltagecommand computed by the vector control device exceeds the maximumvoltage that the inverter can output, a difference between the inverteroutput voltage command and the voltage that the inverter can actuallyoutput is inputted to a magnetic flux correction controller, so that thesecondary magnetic flux command is lowered by an output of the magneticflux correction controller.

Non-Patent Document 1: “Denatsu kotei moudo deno yuuden dendouki nobekutoru seigyo”, Journal of IEEJ, Vol. 118-D, No. 9, 1998.

DISCLOSURE OF THE INVENTION Problems that the Invention is to Solve

However, according to the vector control method of an induction motordisclosed in Non-Patent Document 1, after the inverter output voltagecommand deviates from the voltage that the inverter can actually output,the magnetic flux correction controller operates to adjust the secondarymagnetic reflux command so that the inverter output voltage command islowered, and operates to bring the inverter output voltage command intocoincidence with the maximum voltage that the inverter can actuallyoutput.

In short, the vector control of an induction motor disclosed inNon-Patent Document 1 is configured to correct the inverter outputvoltage command by so-called a feedback loop.

Accordingly, there is a discrepancy between the inverter output voltagecommand and the output voltage of the inverter until the inverter outputvoltage command is corrected appropriately, which raises a problem thatstable vector control cannot be performed.

In addition, it is necessary to add a feedback loop and to add amagnetic flux correction controller as a component of the feedback loop.It is therefore necessary to design control constants, which raisesanother problem that time and labor are required.

The invention is devised to solve the problems as discussed above andhas an object to provide a vector control device of an induction motor,a vector control method of an induction motor, and a drive controldevice of an induction motor capable of performing stable vector controlover the entire range from a low speed range to a high speed range ofthe induction motor without using the feedback loop.

Means for Solving the Problems

A vector control device of an induction motor of the invention is avector control device that controls driving of an induction motor via aninverter, including: secondary magnetic flux command computing means forcomputing a secondary magnetic flux command to the induction motor bytaking a maximum voltage that the inverter can generate into account ona basis of a torque command from an external, a DC voltage to beinputted into the inverter, and an inverter angular frequency, which isan angular frequency of an AC voltage to be outputted from the inverter;q-axis/d-axis current command generating means for generating a q-axiscurrent command and a d-axis current command on a d-q axes rotatingcoordinate system in reference to a secondary magnetic flux of theinduction motor on a basis of the torque command and the secondarymagnetic flux command; output voltage computing means for computing anoutput voltage that the inverter is to output on a basis of the q-axiscurrent command, the d-axis current command, and a circuit constant ofthe induction motor; and voltage command/PWM signal generating means forcontrolling the inverter for the inverter to output the output voltage.

Also, a vector control method of an induction motor of the invention isa vector control method of controlling driving of an induction motor viaan inverter, including: computing a secondary magnetic flux command tothe induction motor by taking a maximum voltage that the inverter cangenerate into account on a basis of a torque command from an external, aDC voltage to be inputted into the inverter, and an inverter angularfrequency, which is an angular frequency of an AC voltage to beoutputted from the inverter; generating a q-axis current command and ad-axis current command on a d-q axes rotating coordinate system inreference to a secondary magnetic flux of the induction motor on a basisof the torque command and the secondary magnetic flux command; computingan output voltage that the inverter is to output on a basis of theq-axis current command, the d-axis current command, and a circuitconstant of the induction motor; and controlling the inverter for theinverter to output the output voltage.

A drive control device of an induction motor of the invention includes:an inverter configured to control driving of an induction motor;secondary magnetic flux command computing means for computing asecondary magnetic flux command to the induction motor by taking amaximum voltage that the inverter can generate into account on a basisof a torque command from an external, a DC voltage to be inputted intothe inverter, and an inverter angular frequency, which is an angularfrequency of an AC voltage to be outputted from the inverter;q-axis/d-axis current command generating means for generating a q-axiscurrent command and a d-axis current command on a d-q axes rotatingcoordinate system in reference to a secondary magnetic flux of theinduction motor on a basis of the torque command and the secondarymagnetic flux command; output voltage computing means for computing anoutput voltage that the inverter is to output on a basis of the q-axiscurrent command, the d-axis current command, and a circuit constant ofthe induction motor; and voltage command/PWM signal generating means forcontrolling the inverter for the inverter to output the output voltage.

Effects of the Invention

According to the invention, the secondary magnetic flux command to theinduction motor is generated in a feed forward manner independently ofthe output voltage saturation state of the inverter. It is thus possibleto perform stable vector control over the entire range from a low speedrange to a high speed range of an induction motor without using afeedback loop for generating the secondary magnetic flux command.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the configuration of a vector controldevice of an induction motor according to a first embodiment of theinvention.

FIG. 2 is a block diagram showing an example of the configuration of asecondary magnetic flux command computation portion in the firstembodiment.

FIG. 3 is a view used to describe behaviors of an internal signal of thesecondary magnetic flux command computation portion in the firstembodiment.

FIG. 4 is a block diagram showing an example of the configuration of avoltage command/PWM signal generation portion in the first embodiment.

FIG. 5 is a view used to describe operations of the vector controldevice of an induction motor in the first embodiment.

FIG. 6 is a view showing a simulation waveform in the first embodiment.

FIG. 7 is a view showing a torque response simulation waveform in thefirst embodiment.

DESCRIPTION OF REFERENCE NUMERALS AND SIGNS

-   1: DC power supply-   2: reactor-   3: capacitor-   4: inverter-   5 a through 5 c: current detectors-   6: motor-   7: speed detector-   8: q-axis current command generation portion-   9: d-axis current command generation portion-   10 and 11: subtracters-   12: g-axis current controller-   13: d-axis current controller-   14: voltage non-interference computation portion-   17 and 18: adders-   19: slip angular frequency command generation portion-   20: secondary resistance correction portion-   21: adder-   22: integrator-   23: three-phase/d-q coordinate transformer-   40: secondary magnetic flux command computation portion-   41: output voltage maximum value computation portion-   42: maximum voltage secondary magnetic flux command computation    portion-   43: switch-   44: lower-order preference portion-   50: voltage command/PWM signal generation portion-   51: modulation index computation portion-   52: voltage phase angle computation portion-   53: multiplier-   54: adjustment gain table-   55: voltage command computation portion-   56: adder-   57: multi-pulse carrier signal generation portion-   58: synchronous three-pulse carrier signal generation portion-   59: switch-   60: pulse mode switching processing portion-   61 through 63: comparators-   64 through 66: inverting circuits-   100: vector control device

Best Mode for Carrying Out the Invention

Hereinafter, one embodiment of the invention will be described on thebasis of the drawings.

The same reference numerals and sings in the respective drawings denotethe same or equivalent components.

First Embodiment

FIG. 1 is a block diagram showing an example of the configuration of avector control device of an induction motor according to a firstembodiment of the invention.

As is shown in the drawing, a main circuit has a DC power supply 1, anLC filter circuit formed of a reactor 2 and a capacitor 3 to suppress aharmonic current from flowing to the power supply side, an inverter 4that converts a DC voltage Efc of the capacitor 3 to an AC voltage at anarbitrary frequency, and a vector control device 100 that performsvector control on an induction motor (hereinafter, referred to simply asthe motor) 6.

It may be thought that the inverter 4 and the vector control device 100together constitute a drive control device that controls the driving ofthe motor 6 by vector control.

The vector control device 100 is configured in such a manner that asignal from a speed detector 7 that detects a rotating speed of themotor 6, signals from current detectors 5 a through 5 c that detectcurrents, the voltage Efc of the capacitor 3 (more specifically, a DCvoltage that is a voltage to be applied from the DC power supply 1 tothe inverter 4 after it is smoothed by the capacitor 3) are inputtedtherein and also a torque command Tm* from an unillustrated externalcontrol device (for example, a system control portion) is inputtedtherein, thereby controlling a torque Tm generated by the motor 6 tocoincide with the torque command Tm*.

By providing the current detectors for at least two phases, the currentof the remaining one phase can be calculated through computation.

In addition, “speed sensor-less vector control method” by which therotating speed of the motor 6 is calculated through computation withoutproviding the speed detector 7 is now put into practical use. In such acase, the speed detector 7 is omitted.

The vector control device 100 controls the motor on the d-q axesrotating coordinate system by defining an axis coinciding with thesecondary magnetic flux axis of the motor 6 as the d-axis and an axisorthogonal to the d-axis as the q-axis, and is configured to performso-called vector control.

Hereinafter, the configurations and the operations of the respectivecomponents forming the vector control device 100 will be described.

As is shown in FIG. 1, a q-axis current command generation portion 8 anda d-axis current command generation portion 9 compute respectively ad-axis (excitation) current command Id* and a q-axis (torque) currentcommand Iq*, respectively, in accordance with Equations (1) and (2)below using the torque command Tm* inputted from the external controldevice (not shown), a secondary magnetic flux command φ2* generated bythe secondary magnetic flux computation portion 40, and circuitconstants of the motor 6:Iq*=(Tm*/(φ2*·PP))·(L2/M)  (1)Id*=φ2*/M+L2/(M·R2)·sφ2*  (2).

Herein, in Equations (1) and (2) above, L2 is a secondaryself-inductance of the motor and expressed as L2=M+l2. Also, M is amutual inductance, l2 is a secondary leakage inductance, s is adifferential operator, PP is pairs of poles of the motor 6, and R2 issecondary resistance of the motor 6.

The secondary magnetic flux command computing portion 40 is the portionforming the centerpiece of the invention and the detailed configurationand the operation will be described below.

Subsequently, a slip angular frequency command generation portion 19computes a slip angular frequency command ωs* to be provided to themotor 6 in accordance with Equation (3) below using the d-axis currentcommand Id*, the q-axis current command Iq*, and the circuit constantsof the motor 6:ωs*=(Iq*/Id*)·(R2/L2)  (3).

Herein, in Equation (3) above, R2 is secondary resistance of the motor.

A secondary resistance correction portion 20 is configured to obtain asecondary resistance correcting value PFS in accordance with Equation(4) below by performing proportional-plus-integral control on adifference between the q-axis current command Iq* and the q-axis currentIq.

This configuration aims at compensating for, of the constants of themotor 6, “a change of the secondary resistance R2 with temperature” thatgives significant influences to the torque control performance.

The secondary resistance correcting value PFS is outputted in accordancewith Equation (4) below only in a control mode 2 described below and itis set to 0 in a control mode 1 described below.PFS=(K3+K4/s)·(Iq*−Iq)  (4)

Herein, in Equation (4) above, s is a differential operator, K3 is aproportional gain, and K4 is an integral gain. The proportional gain K3is a coefficient to multiply a deviation between Iq* and Iq and theintegral gain K4 is a coefficient to multiply an integral term of thedeviation between Iq* and Iq.

The slip angular frequency command ωs* calculated in accordance withEquation (3) above, a rotating angular frequency ωr as an output of thespeed detector 7 attached to the axial end of the motor 6, and thesecondary resistance correcting value PFS as an output of the secondaryresistance correction portion 20 are added by an adder 21 and let thesum be an inverter angular frequency ω to be outputted from the inverter4. Then, the inverter angular frequency ω is integrated by an integrator22, and the result of integration is inputted into a voltage command/PWMsignal generation portion 50 and a three-phase/d-q axes coordinatetransformer 23 described below as the basic phase angle θ of thecoordinate transformation.

The three-phase/d-q axes coordinate transformer 23 converts a U-phasecurrent Iu, a V-phase current Iv, a W-phase current Iw detected by thecurrent detectors 5 a through 5 c, respectively, to a d-axis current Idand a q-axis current Iq on the d-q coordinate calculated in accordancewith Equation (5) below.

$\begin{matrix}{\begin{pmatrix}{Iq} \\{Id}\end{pmatrix} = {\sqrt{\frac{2}{3}}{\begin{pmatrix}{\cos\;\theta} & {\cos\left( {\theta - {\frac{2}{3}\pi}} \right)} & {\cos\left( {\theta + {\frac{2}{3}\pi}} \right)} \\{\sin\;\theta} & {\sin\left( {\theta - {\frac{2}{3}\pi}} \right)} & {\sin\left( {\theta + {\frac{2}{3}\pi}} \right)}\end{pmatrix} \cdot \begin{pmatrix}{IU} \\{IV} \\{IW}\end{pmatrix}}}} & (5)\end{matrix}$

Subsequently, a subtracter 10 finds a difference between the q-axiscurrent command Iq* and the q-axis current Iq, and inputs the result(that is, the difference between Iq* and Iq) into a q-axis currentcontroller 12 in the next stage.

The q-axis current controller 12 performs proportional-plus-integralcontrol on the input value (that is, the difference between Iq* and Iq)and outputs a q-axis voltage compensating value qe.

Also, another subtracter 11 finds a difference between the d-axiscurrent command Id* and the d-axis current Id, and inputs the result(that is, the difference between Id* and Id) into a d-axis currentcontroller 13 in the next stage.

The d-axis current controller 13 performs proportional-plus-integralcontrol on the input value (that is, the difference between Id* and Id),and outputs a d-axis voltage compensating value de.

The q-axis compensating value qe and the d-axis compensating value deare expressed, respectively, by Equations (6) and (7) below:qe=(K1+K2/s)·(Iq*−Iq)  (6)de=(K1+K2/s)·(Id*−Id)  (7).

Herein, in Equations (6) and (7) above, s is a differential operator, K1is a proportional gain, and K2 is an integral gain.

As will be described below, after the control mode 1 (described below)has shifted to the control mode 2 (described below), qe and de aregradually reduced to 0.

Subsequently, a voltage non-interference computation portion 14 computesa d-axis feed forward voltage Ed* and a q-axis feed forward voltage Eq*,respectively, in accordance with Equations (8) and (9) below using thed-axis current command Id*, the q-axis current command Iq*, and thecircuit constants of the motor 6:

$\begin{matrix}{{Ed}^{*} = {{\left( {{R\; 1} + {{s \cdot L}\;{1 \cdot \sigma}}} \right) \cdot {Id}^{*}} - {{\omega \cdot L}\;{1 \cdot \sigma \cdot {Iq}^{*}}} + {{\left( {{M/L}\; 2} \right) \cdot s}\;\phi\; 2^{*}}}} & (8) \\{{Eq}^{*} = {{\left( {{R\; 1} + {{s \cdot L}\;{1 \cdot \sigma}}} \right) \cdot {Iq}^{*}} + {{\omega \cdot L}\;{1 \cdot \sigma \cdot {Id}^{*}}} + {{\left( {{\omega \cdot M \cdot \phi}\; 2^{*}} \right)/L}\; 2.}}} & (9)\end{matrix}$

Herein, in Equations (8) and (9) above, σ is a leakage coefficientdefined as σ=1−M²/(L1·L2).

Also, R1 is primary resistance of the motor 6 and L1 is a primaryself-inductance of the motor 6 calculated as L1=M+l1.

L2 is a secondary self-inductance of the motor 6 calculated as L2=M+l2.

Herein, l1 is a primary leakage inductance and l2 is a secondary leakageinductance.

Ed* and Eq* expressed, respectively, by Equations (8) and (9) above aremade up of the motor constants and the current commands (Iq* and Id*)both are known in advance and include no feedback elements. Hence, theyare referred to as feed forward voltages.

Subsequently, the q-axis voltage compensating value qe and the q-axisfeed forward voltage Eq* are added by an adder 17 and the d-axis voltagecompensating value de and the d-axis feed forward voltage Ed* are addedby another adder 18. The sum of the former and the sum of the latter areinputted into the voltage command/PWM signal generation portion 50 as aq-axis voltage command Vq* and a d-axis voltage command Vd*,respectively.

The q-axis voltage command Vq* and the d-axis voltage command Vd* areexpressed, respectively, by Equations (10) and (11) below:Vq*=Eq*+qe  (10)Vd*=Ed*+de  (11).

An inverter output voltage command VM* in this instance is expressed byEquation (12) below:VM*=(Vd* ² +Vq* ²)^(1/2)  (12)

Herein, VM* represents the magnitude of an inverter output voltagecommand vector.

It should be noted that the voltage non-interference computation portion14 and the adders 17 and 18 together constitute output voltage computingmeans for computing the output voltage that the inverter 4 is to output.

Finally, gate signals to the switching elements U through Z (not shown)of the invert 4 are outputted from the voltage command/PWM signalgeneration portion 50.

Because the inverter 4 is a known voltage-source PWM inverter, thedetailed configuration is omitted herein. However, to add somedescription in part, the switching elements U, V, and W are theswitching elements disposed, respectively, in the U-phase, the V-phase,and the W-phase on the upper arm of the inverter 4, and the switchingelements X, Y, and Z are switching elements disposed, respectively, inthe U-phase, the V-phase, and the W-phase on the lower arm of theinverter 4.

The configurations of the secondary magnetic flux command computationportion 40 and the voltage command/PWM signal generation portion 50,which are important components of the invention, will now be described.

FIG. 2 is a view showing an example of the configuration of thesecondary magnetic flux command computation portion 40 of thisembodiment.

As is shown in FIG. 2, the capacitor voltage Efc, the torque commandTm*, the inverter angular frequency ω, a powering secondary magneticflux command φ2P*, and a brake secondary magnetic flux command φ2B* areinputted into the secondary magnetic flux command computation portion40.

The output voltage maximum computation portion 41 calculates the maximumvalue VMmax of the inverter output voltage VM in accordance withEquation (13) below using the capacitor voltage Efc.

$\begin{matrix}{{VMmax} = {\frac{\sqrt{6}}{\pi} \cdot {Efc}}} & (13)\end{matrix}$

Herein, VMmax is the maximum voltage that the inverter can output on thecapacitor voltage Efc, and it is a value when the inverter 4 is operatedin a, single-pulse mode in which the output line-to-line voltagewaveform is of 120° rectangular-wave conduction.

Equation (13) above is an equation disclosed also in Non-Patent Document1 specified above, and it is obtained as the fundamental wave componentwhen the rectangular wave of 120° conduction is expanded by a Fourierseries.

A secondary magnetic flux command φ2H* that is needed exactly to bringthe inverter output voltage VM into coincidence with the maximum valueVMmax is calculated by a maximum voltage secondary magnetic flux commandcomputation portion 42 in accordance with Equation (14) below using themaximum value VMmax of the inverter output voltage VM calculated inaccordance with Equation (13) above, the torque command Tm*, theinverter angular frequency ω, and the constants of the motor 6.

$\begin{matrix}{{\Phi\; 2H^{*}} = \sqrt{\frac{{- A} + \sqrt{A^{2} - B}}{C}}} & (14)\end{matrix}$where we define

A = 2 ⋅ R 1 ⋅ ω ⋅ Tm^(*) − VMmax²$B = {{4 \cdot \frac{\left\{ {{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}} \right\} \cdot \left\{ {{{RI}\;}^{2} + {\sigma^{2}\left( {{\omega \cdot L}\; 1} \right)}^{2}} \right\}}{M^{4}} \cdot {Tm}^{*^{2}} \cdot L}\; 2^{2}}$$C = {2 \cdot \frac{{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}}{M^{2}}}$

Because Equation (14) above is an important equation to constitute theinvention, the derivation process will be described briefly below.

On the condition that a time change of the d-axis secondary magneticflux is moderate, a transient term is neglected from the circuitequation (known) of the motor 6 in a state where vector control isestablished on the d-q axes, then a d-axis voltage Vd of the motor 6 canbe obtained in accordance with Equation (15) below and a q-axis voltageVq of the motor 6 can be obtained in accordance with Equation (16)below:Vd=R1·Id−ω·L1·σ·Iq  (15)Vq=R1·Iq+ω·L1·σ·Id+(ω·M·φ2*)/L2  (16)where Vd is the d-axis voltage of the motor 6 and Vq is the q-axisvoltage of the motor 6.

In addition, we find Equation (17) below from the circuit equation(known) of the motor 6:−M·R2·Id+(R2+s·L2)·φ2=0  (17).

Herein, in Equation (17) above, φ2 is the d-axis secondary magnetic fluxof the motor 6.

Herein, by neglecting the transient term of Equation (17) above on thecondition that a change of the d-axis secondary magnetic flux φ2 ismoderate, we find Equation (18) below, which is a relational expressionof the d-axis current Id and the d-axis secondary magnetic flux φ2:Id=φ2/M  (18).

In a case where vector control is established, we find Equation (19)below (known), which is a relational expression of the q-axis current Iqand the torque Tm:Tm=(M/L2)·Iq·φ2  (19).

By modifying Equation (19) above, we find Equation (20) below:Iq=(Tm·L2)/M ²  (20).

By substituting Equation (18) above, which is the relational expressionof the d-axis current Id and the d-axis secondary magnetic flux φ2, andEquation (20) above, which is the relational expression of the q-axiscurrent Iq and the torque Tm, into Equation (15) and Equation (16), wefind Equation (21) and Equation (22) below as the d-q axes voltage ofthe motor 6:

$\begin{matrix}{{Vd} = {{R\;{1 \cdot \left( {\phi\;{2/M}} \right)}} - {{\omega \cdot L}\;{1 \cdot \sigma \cdot {\left( {{{Tm}/L}\; 2} \right)/\left( {\phi\;{2 \cdot M}} \right)}}}}} & (21) \\{{Vq} = {{R\;{1 \cdot {\left( {{{Tm} \cdot L}\; 2} \right)/\left( {\phi\;{2 \cdot M}} \right)}}} + {{\omega \cdot \phi}\;{2 \cdot L}\;{1/M}}}} & (22)\end{matrix}$

Herein, let VM² be the value of a sum of the square of Equation (21)above and the square of Equation (22) above, then we find Equation (23)below.

$\begin{matrix}\begin{matrix}{{VM}^{2} = {{Vd}^{2} + {Vq}^{2}}} \\{= {{{\frac{R\; 1^{2}\left( {{\omega \cdot L}\; 1} \right)^{2}}{M} \cdot \Phi}\; 2} + {\frac{R\; 1^{2}\left( {{\omega \cdot L}\;{1 \cdot \sigma}} \right)^{2}}{{M^{2} \cdot \Phi}\; 2^{2}}\left( {{Tm} \cdot {L2}} \right)^{2}} +}} \\{{2 \cdot R}\;{1 \cdot \frac{{\omega \cdot L}\;{1 \cdot L}\;{2 \cdot {Tm}}}{M^{2}}}\left( {1 - \sigma} \right)}\end{matrix} & (23)\end{matrix}$

It should be noted that VM is the voltage of the motor 6 and because thevoltage of the motor 6 is equal to an output voltage of the inverter 4,the term, “the inverter output voltage VM”, is used in the followingdescriptions.

By multiplying the both sides of Equation (23) above by φ2² forarrangement, we find a quadratic equation with respect to the d-axissecondary magnetic flux φ2 of the motor 6.

By finding the solution, we find Equation (24) below.

$\begin{matrix}{{\Phi\; 2} = \sqrt{\frac{{- D} + \sqrt{D^{2} - E}}{F}}} & (24)\end{matrix}$where we define

D = 2 ⋅ R 1 ⋅ ω ⋅ Tm − VM²$E = {{4 \cdot \frac{\left\{ {{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}} \right\} \cdot \left\{ {{R\; 1^{2}} + {\sigma^{2}\left( {{\omega \cdot L}\; 1} \right)}^{2}} \right\}}{M^{4}} \cdot T}\;{m^{2} \cdot L}\; 2^{2}}$$F = {2 \cdot \frac{{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}}{M^{2}}}$

It is understood that Equation (24) expresses the relation among thed-axis secondary magnetic flux φ2 of the motor 6, the inverter outputvoltage VM, the inverter angular frequency ω, the torque Tm of the motor6, and the constants (R1, L1, L2, and M) of the motor 6.

By substituting the maximum value VMmax as the inverter output voltageVM, Equation (24) above expresses the relation among a generation torqueTm of the motor 6 at VMmax, the d-axis secondary magnetic flux φ2, andthe inverter angular frequency ω.

In order to apply this relation at the control end, by replacing thed-axis secondary magnetic flux φ2 in Equation (24) above with themaximum voltage secondary magnetic flux command φ2H* and by replacingthe torque Tm with the torque command Tm*, we find Equation (14) above.

As can be understood from the foregoing, the maximum voltage secondarymagnetic flux command φ2H* obtained in accordance with Equation (14)above is the secondary magnetic flux command that is needed exactly tobring the inverter output voltage VM into coincidence with the maximumvalue VMmax that the inverter can output under the condition that themotor 6 is run by the torque command Tm* at the inverter angularfrequency ω.

In other words, the inverter output voltage command VM* computed by thevector control device 100 using the maximum voltage secondary magneticflux command φ2H* takes the value that is needed exactly to bring theinverter output voltage VM into coincidence with the maximum value VMmaxthat the inverter can output, and the inverter output voltage commandVM* will never deviate from the maximum value VMmax that the invertercan output.

It is general to apply a certain rated secondary magnetic flux to themotor 6 since the motor 6 is activated until the output voltage of theinverter saturates.

It is general to ensure the rated secondary magnetic flux to the largestextent possible under the condition that the central core of iron of themotor 6 will not undergo magnetic saturation.

The optimal value differs during powering and during regeneration of themotor 6. Accordingly, as is shown in FIG. 2, a powering rated secondarymagnetic flux command φ2P* is used during powering and a regenerationrated secondary magnetic flux command φ2B* is used during regenerationby switching from one to the other with a switch 43, and an output ofthe switch 43 is defined as a rated secondary magnetic flux commandφ2C*.

The powering rated secondary magnetic flux command φ2P* and theregeneration rated secondary magnetic flux command φ2B* may bedetermined arbitrarily under the conditions specified above. However,they can be computed on off-line by substituting the maximum value VMmaxof the inverter output voltage VM calculated by substituting a nominalDC voltage (for example, 1500 V for a typical railway) for Efc inEquation (13) above, the rated value of the torque command Tm*, theinverter angular frequency ω equal to the base frequency of the motorregulated by the vehicle performance of an electric vehicle, and theconstants of the motor 6 into Equation (14) above, so that they are setpreliminary in the vector control device 100. When configured in thismanner, it becomes easier to design the constants of the vector controldevice 100.

Subsequently, a lower-order preference portion 44 chooses either themaximum voltage secondary magnetic flux command φ2H* or the ratedsecondary magnetic flux command φ2C*, whichever is the smaller, andgenerates a secondary magnetic flux command φ2* to be used ultimatelyfor vector control.

Behaviors of an internal signal of the secondary magnetic flux commandcomputation portion 40 configured as above will be described below.

FIG. 3 is a view used to describe behaviors of an internal signal of thesecondary magnetic flux command computation portion 40 according to thisembodiment of the invention.

As is shown in FIG. 3, as the secondary magnetic flux command φ2* usedfor vector control, the rated secondary magnetic flux command φ2C* ischosen until the output voltage of the inverter saturates (the range onthe left from a capital S in FIG. 3) and the maximum voltage secondarymagnetic flux command φ2H* is chosen in the output voltage saturationrange of the inverter (the range on the right of the capital S in FIG.3).

Owing to these operations, it is possible to obtain the secondarymagnetic flux command φ2* that is needed exactly to bring the inverteroutput voltage VM into coincidence with the maximum value VMmax in theoutput voltage saturation range of the inverter in real time.

That is to say, because the secondary magnetic flux command φ2* isdetermined instantaneously without any time delay in accordance with thecomputing equation expressed by Equation (14) above having no feedbackelement using the motor constants and known amounts, the necessarysecondary magnetic flux command φ2* can be obtained in real time in afeed forward manner.

The configuration of the voltage command/PWM signal generation portion50 will now be described.

FIG. 4 is a view showing an example of the configuration of the voltagecommand/PWM signal generation portion 50 of this embodiment.

As is shown in FIG. 4, a modulation index computation portion 51 and avoltage phase angle computation portion 52 calculate percent modulationPMF and a voltage phase angle THV, respectively, using the inverteroutput voltage command VM* expressed by Equation (12) above, the maximumvalue VMmax of the inverter output voltage VM expressed by Equation (13)above, the d-axis voltage command Vd*, and the q-axis voltage commandVq*.

The modulation index computation portion 51 and the voltage phase anglecomputation portion 52 respectively compute Equations (25) and (26)below.

$\begin{matrix}{{PMF} = \frac{{VM}^{*}}{VMmax}} & (25) \\{{THV} = {\tan^{- 1} \cdot \frac{{Vq}^{*}}{{Vd}^{*}}}} & (26)\end{matrix}$

The voltage phase angle THV is added to the base phase angle θ by anadder 56 and the sum is inputted into a voltage command computationportion 55 and a synchronous three-pulse carrier signal generationportion 58 as a control phase angle θ1.

The percent modulation PMF represents a ratio of the inverter outputvoltage command VM* with respect to the maximum voltage VMmax (definedby Equation (13) above) that the inverter can output. It indicates thatthe inverter output voltage command VM* becomes equal to the maximumvalue VMmax of the inverter output voltage in a case where PMF=1.0.

The value found by multiplying the percent modulation PMF by an outputof an adjustment gain table 54 by a multiplier 53 is inputted into thevoltage command computation portion 55 as voltage command amplitudePMFM.

The adjustment gain table 54 is to correct a variance of the relation ofthe inverter output voltage VM with respect to the percent modulationPMF in the multi-pulse PWM mode and the synchronous three-pulse PWM modeand the summary is as follows.

The maximum voltage (effective value) that the inverter 4 can outputwithout any distortion is 0.612·Efc in the multi-pulse PWM mode and0.7797·Efc in the synchronous three-pulse PWM mode.

In short, the output voltage of the inverter with respect the percentmodulation PMF in the multi-pulse PWM mode is 1/1.274 of that in thesynchronous three-pulse PWM mode.

In order to cancel out this difference, the percent modulation PMF isincreased by 1.274 times in the multi-pulse PWM mode and then inputtedinto the voltage command computation portion 55 as the voltage commandamplitude PMFM.

The voltage command computation portion 55 generates a U-phase voltagecommand Vu*, a V-phase voltage command Vv*, and a W-phase voltagecommand Vw* in accordance with computing equations expressed byEquations (27) through (29) below, respectively, using the percentmodulation PMF and the control phase angle θ1.

$\begin{matrix}{{Vu}^{*} = {{{PMFM} \cdot \sin}\;\theta\; 1}} & (27) \\{{Vv}^{*} = {{{PMFM} \cdot \sin}\;\left( {{\theta\; 1} - \frac{2\;\pi}{3}} \right)}} & (28) \\{{Vw}^{*} = {{{PMFM} \cdot \sin}\;\left( {{\theta\; 1} - \frac{4\;\pi}{3}} \right)}} & (29)\end{matrix}$

The U-phase voltage command Vu*, the V-phase voltage command Vv*, andthe W-phase voltage command Vw* are compared with a carrier signal CARin magnitude by comparators 61 through 63, respectively, and gatesignals U, V, and W are generated while gate signals X, Y, and Z aregenerated via inverting circuits 64 through 66, respectively.

The carrier signal CAR is a signal chosen by a pulse mode switchingprocessing portion 60 by means of a switch 59 from a multi-pulse(generally, in the neighborhood of 1 kHz) carrier signal A generated bya multi-pulse carrier signal generation portion 57, a synchronousthree-pulse carrier signal B generated by a synchronous three-pulsecarrier signal generation portion 58, and a zero value C chosen in thesingle-pulse mode.

The pulse mode switching processing portion 60 operates to cause theswitch 59 to switch to an asynchronous carrier A side in a range wherethe percent modulation PMF is low (0.785 or lower), to a synchronousthree-pulse carrier B side in a range where the percent modulation PMFis 0.785 to 1.0 both exclusive, and to the zero value C side when thepercent modulation PMF reaches 1.0, depending on the percent modulationPMF and the control phase angle θ1.

By configuring in this manner, it is possible to switch the pulse modeto the single-pulse mode at the same timing at which the percentmodulation PMF reaches 1.0, that is, the inverter output voltage VMbecomes equal to the maximum value VMmax.

Each of computing equations specified above are generally carried out byS/W processing in a microcomputer. In a case where the computationaccuracy (the number of bits) is reduced with the aim of reducing thecomputation load on the microcomputer or any other reasonable aim, thepercent modulation PMF does not reach exactly 1.0 at the timing at whichthe inverter output voltage VM becomes equal to the maximum value VMmaxand may possibly take a smaller value, for example, 0.999 . . . .

However, in this case, too, the invention is feasible when the percentmodulation PMF is 0.95 or higher, although a minor voltage jump occurseven when the pulse mode is switched to the single-pulse mode.

FIG. 5 is a view used to describe transition of the inverter angularfrequency ω, the percent modulation PMF, and the pulse mode, operationsof the switch 59 to switch the control pulse mode, and transition of thecontrol mode in this embodiment.

As is shown in FIG. 5, when an electric vehicle is at a low speed, thatis, when the inverter angular frequency ω is low, the percent modulationPMF is small and the pulse mode is the multi-pulse PWM mode and theswitch 59 chooses A (see FIG. 4).

Also, the control mode is the control mode land the q-axis currentcontroller 12 and the d-axis current controller 13 operate in accordancewith Equations (6) and (7) above, respectively.

When the speed of the electric vehicle increases and the percentmodulation PMF reaches or exceeds 0.785, because the output voltagesaturates in the multi-pulse PWM mode, the switch 59 is switched to Band the pulse mode is switched to the synchronous three-pulse PWM mode.

Herein, the synchronous three-pulse mode is a mode necessary to output avoltage at the percent modulation PMF of 0.785 or higher.

In the multi-pulse PWM mode, it is impossible to output a voltage at thepercent modulation PMF of 0.785 or higher unless over modulation (knownart) is employed.

In addition, the control mode 2 is chosen as the control mode and theq-axis current controller 12 and the d-axis current controller 13 stopcomputations and the outputs are reduced to 0.

The outputs are reduced to 0 for the reason as follows. That is, becausethe number of pulses in the inverter output voltage half cycle in thesynchronous three-pulse PWM mode is reduced to three from ten or more inthe multi-pulse PWM mode, the control delay increases, and whencomputations by the q-axis current controller 12 and the d-axis currentcontroller 13 continue in this state, there is a risk that thesecontrollers become unstable. The computations of the q-axis currentcontroller 12 and the d-axis current controller 13 are stopped to avoidsuch a risk.

In the control mode 2, the secondary resistance correction portion 20starts to operate and computes the secondary resistance correcting valuePFS in accordance with Equation (4) above.

When the speed of the electric vehicle increases further and the percentmodulation PMF reaches 1.0, the switch 59 is switched to C and the pulsemode is switched to the single-pulse mode. The control mode remains inthe control mode 2.

A case where the electric vehicle decreases the speed by putting on theregenerative brake is not shown in the drawing. However, the pulse modeis switched from the single-pulse mode to the synchronous three-pulsePWM mode to the multi-pulse PWM mode, the switch 59 switches from C to Bto A (see FIG. 4), and the control mode shifts from the control mode 2to the control mode 1 in the order inverse to the order described above.

FIG. 6 is a view showing a simulation waveform of this embodiment.

FIG. 6 shows a case where the motor 6 is accelerated by powering bylaunching the torque command Tm* at the time about 0.8 (s) under thecondition, capacitor voltage Efc=1500 V.

The multi-pulse PWM mode and the control mode 1 are chosen in aninterval from the times about 0.8 (s) to 3.5 (s), and the ratedsecondary magnetic flux φ2C* is chosen as the secondary magnetic fluxcommand φ2*. The motor 6 is thus excited by a certain magnetic flux.

Accordingly, the q-axis voltage command Vq* and the d-axis voltagecommand Vd* increase in magnitude in proportion to acceleration of themotor and so does the inverter output voltage command VM*. The percentmodulation PMF also increases in association with the increasinginverter output voltage command VM*, which causes the U-phase voltagecommand Vu* to increase. The torque Tm of the motor 6 accelerates byfollowing Tm* in a stable manner.

Subsequently, the pulse mode is switched to the synchronous three-pulsemode at the time about 3.5 (s) and the control mode is switched to thecontrol mode 2.

The secondary magnetic flux command φ2* remains as the rated secondarymagnetic flux φ2C* and the motor 6 is excited by a certain magneticflux.

Accordingly, the q-axis voltage command Vq* and the d-axis voltagecommand Vd* continue to increase in magnitude in proportion toacceleration of the motor 6 and so does the inverter output voltagecommand VM*. The percent modulation PMF increases in association withthe increasing inverter output voltage command VM*, which causes theU-phase voltage command Vu* to increase.

The amplitude of the U-phase voltage command Vu* reduces immediatelyafter the switching to the synchronous three-pulse PWM mode. This isbecause the voltage command amplitude PMFM that has been increased by1.274 times by the adjustment gain table 54 in the multi-pulse PWM modeas described above is switched and the scale factor is set to 1.0.

The torque Tm of the motor 6 accelerates by following Tm* in a stablemanner.

Ripples are observed in the torque Tm for a while since the time about3.5 (s). This is because the number of pulses is so small in thesynchronous three-pulse PWM mode that a current ripple of the motor 6increases. However, such ripples are negligible when an electric vehiclehaving a large inertia is driven. The mean value of the torque Tmcoincides with the torque command Tm* and the torque Tm is thereforecontrolled in a stable manner.

Subsequently, the inverter output voltage saturates at the time about4.6 (s) and at the same time the maximum voltage secondary magnetic fluxcommand φ2H* computed in accordance with Equation (14) above is chosenas the secondary magnetic flux command φ2* by the secondary magneticflux command computation portion 40 (see FIG. 1).

Accordingly, the percent modulation PMF is fixed at 1.0 and the inverteroutput voltage command VM* is fixed to the maximum voltage VMmax thatthe inverter can output (in this case, VMmax is found to be about 1170 Vby substituting Efc=1500 V into Equation (13) above).

The torque command Tm* is reduced in inversely proportional to therotating number in order to run the motor 6 with a constant output. Itis, however, understood that the torque Tm of the motor 6 accelerates ina stable manner by following Tm*.

FIG. 7 is a view showing a torque response simulation waveform in thisembodiment.

FIG. 7 is a response waveform of the torque Tm of the motor 6 when thetorque command Tm* is decreased and increased stepwise in a single-pulsemode range (an interval from the times 5.3 (s) to 5.9 (s)) of FIG. 6.

It is understood that, as is shown in FIG. 7, a high-speed response atthe time constant of 10 ms or smaller is obtained and high-speed torquecontrol by vector control is achieved even in the single-pulse mode inthe voltage saturation range of the inverter.

Also, even in a case where the capacitor voltage Efc varies, it isobvious from Equation (13) and Equation (14) above that the secondarymagnetic flux command φ2* responding to such a variance is calculated,and the control can be achieved in a stable manner in this case, too.

As has been described, according to this embodiment, it is possible tocalculate the secondary magnetic flux command φ2* that can bring theinverter output voltage command VM* into coincidence with the maximumvoltage VMmax that the inverter can output in accordance with thecomputing equations in real time in a feed forward manner in the voltagesaturation range of the inverter independently of a variance of thetorque command Tm* and the capacitor voltage Efc.

Accordingly, it is possible to achieve a vector control method for thevoltage saturation range capable of, in principle, eliminating an eventthat the inverter output voltage command VM* deviates from the maximumvoltage VMmax that the inverter can output and eliminating the need toset the control constants by making it unnecessary to add a feedbackloop, such as a magnetic flux correction controller.

Further, it is possible to switch the pulse mode to the single-pulsemode at the timing at which the percent modulation PMF reaches 1.0 asthe pulse mode is switched from the multi-pulse PWM mode to thesynchronous three-pulse PWM mode, that is, at the timing at which theinverter output voltage becomes equal to the maximum value VMmax.

It thus becomes possible to obtain a vector control device of aninduction motor capable of performing stable vector control over theentire range from the multi-pulse PWM mode in a low speed range to thesingle-pulse mode at a medium and high speed range, which is the outputvoltage saturation range of the inverter.

The configurations described in the embodiment above are mere examplesof the contents of the invention. It goes without saying that theinvention can be combined with other known techniques and modifiedwithout deviating from the scope of the invention by omitting theconfigurations in part.

In this embodiment, the secondary resistance correction portion thatcorrects the inverter angular frequency from a deviation between theq-axis current command and the q-axis current is operated by operatingthe q-axis current controller and the d-axis current controller in themulti-pulse mode and stopping the q-axis current controller and thed-axis current controller in the mode with three pulses or fewer.

There can be achieved an effect that a secondary magnetic flux commandexceeding the maximum voltage that the inverter can output will not beissued without using feedback control, such as the magnetic fluxcorrection control to find a secondary magnetic flux command byconfiguring in such a manner so as to operate the q-axis currentcontroller and the d-axis current controller independently of the pulsemode, to operate the secondary resistance correction portionindependently of the pulse mode without providing the q-axis currentcontroller and the d-axis current controller, or to provide none of theq-axis current controller, the d-axis current controller, and thesecondary resistance correction portion.

Further, the invention has been described in this specification in viewof a power converting device in the railway field. It should beappreciated, however, that applications of the invention are not limitedto this field. It goes without saying that the invention can be appliedto various related fields, such as an automobile, an elevator, and anelectric power system.

As has been described, a vector control device of an induction motor ofthe invention is a vector control, device that controls driving of aninduction motor (6) via an inverter (4), and includes: secondarymagnetic flux command computing means (40) for computing a secondarymagnetic flux command to the induction motor (6) by taking a maximumvoltage that the inverter (4) can generate into account on a basis of atorque command from an external, a DC voltage to be inputted into theinverter, and an inverter angular frequency, which is an angularfrequency of an AC voltage to be outputted from the inverter;q-axis/d-axis current command generating means (8 and 9) for generatinga q-axis current command and a d-axis current command on a d-q axesrotating coordinate system in reference to a secondary magnetic flux ofthe induction motor (6) on a basis of the torque command and thesecondary magnetic flux command; output voltage computing means (voltagenon-interference computation portion 14, adder 17, and adder 18) forcomputing an output voltage that the inverter (4) is to output on abasis of the q-axis current command, the d-axis current command, and acircuit constant of the induction motor (6); and voltage command/PWMsignal generating means (50) for controlling the inverter (4) for theinverter (4) to output the output voltage.

Accordingly, the secondary magnetic flux command to the induction motoris generated in a feed forward manner independently of the outputvoltage saturation state of the inverter. It is thus possible to performstable vector control over the entire range from a low speed range to ahigh speed range of the induction motor without using a feedback loop.

Also, the secondary magnetic flux command computing means (40) in thevector control device of an inductor motor of the invention has amaximum voltage secondary magnetic flux command computation portion (42)configured to compute a maximum voltage secondary magnetic flux command,which is a secondary magnetic flux command to bring the maximum voltagethat the inverter (4) can generate and the output voltage intocoincidence in magnitude, and a lower-order preference portion (44)configured to output either the maximum voltage secondary magnetic fluxcommand or a pre-set rated secondary magnetic flux command, whichever isthe smaller, as the secondary magnetic flux command.

Accordingly, even in a case where the inverter is in the voltagesaturation range, not only is it possible to generate an inverter outputvoltage command that coincides with the maximum voltage that theinverter can output owing to the maximum voltage secondary magnetic fluxcommand, but it is also possible to automatically switch the ratedsecondary magnetic flux command and the maximum voltage secondarymagnetic flux command in response to the inverter output voltagecommand.

Also, the maximum voltage secondary magnetic flux command computationportion (42) in the vector control device of an induction motor of theinvention computes the maximum voltage secondary magnetic flux commandon a basis of the torque command and the inverter angular frequency.

Because the torque command and the inverter angular frequency are knownand include no feedback elements, it is possible to compute the maximumvoltage secondary magnetic flux command instantaneously with ease.

Also, in the vector control device of an induction motor of theinvention, the maximum voltage secondary magnetic flux command iscomputed in accordance with Equation (14) above.

Because the maximum voltage secondary magnetic flux command isdetermined uniquely in accordance with the computing equation expressedby Equation (14) above including no feedback elements, there is no needto adjust the control constants within the feedback loop and the maximumvoltage secondary magnetic flux command can be computed instantaneouslywith ease in comparison with a case where the feedback loop is included.

Also, in the vector control device of an induction motor of theinvention, the rated secondary magnetic flux command has at least twokinds of values including a value applied during powering of theinduction motor (6) and a value applied during regeneration and isconfigured to be capable of switching the values according to a runningstate of the induction motor (6).

Accordingly, even in a case where the optimal rated secondary magneticflux command for the induction motor differs during powering and duringregeneration, it becomes possible to control the induction motor byapplying the optimal rated secondary magnetic flux command.

Also, in the vector control device of an induction motor of theinvention, the rated secondary magnetic flux command is a value setthrough preliminary computation using the computing equation expressedby Equation (14) above. Accordingly, it is possible to calculate theoptimal rated secondary magnetic flux command easily using the motorconstant.

Also, in the vector control device of an induction motor of theinvention, a pulse mode of the inverter (4) is switched in response topercent modulation of the inverter (4) computed on a basis of thesecondary magnetic flux command and the torque command.

Accordingly, it is possible to change the fundamental wave components ofthe actual output voltage of the inverter continuously according to theinverter output voltage command that varies with the secondary magneticflux command and the inverter frequency.

Also, in the vector control device of an induction motor of theinvention, the inverter (4) is operated in a single-pulse mode whenpercent modulation of the inverter (4) computed on a basis of thesecondary magnetic flux command is 0.95 or higher.

It thus becomes possible to shift the output voltage of the invertercontinuously to the maximum value.

Also, the vector control device of an induction motor of the inventionfurther includes: a current detector (5 a through 5 c) configured tomeasure a current flowing through the induction motor (6); athree-phase/d-q axes coordinate transformer (23) configured to convertthe current detected by the current detector (5 a through 5 c) to aq-axis current and a d-axis current, which are values on the d-q axesrotating coordinate system; q-axis current control means (12) foroperating so as to lessen a deviation between the q-axis current commandand the q-axis current; and d-axis current control means (13) foroperating so as to lessen a deviation between the d-axis current commandand the d-axis current, wherein the output voltage computing means(formed of voltage non-interference computation portion 14, adder 17,and adder 18) computes the output voltage using outputs of the q-axiscurrent control means (12) and the d-axis current control means (13),and computations by the q-axis current control means (12) and the d-axiscurrent control means (13) are stopped in a case where the number ofpulses in a half cycle generated by the inverter (4) is three orsmaller. It is therefore possible to ensure the stability of vectorcontrol.

Also, in the vector control device of an induction motor of theinvention, the inverter angular frequency is corrected using a deviationbetween the q-axis current command and the q-axis current in a casewhere the number of pulses in the half cycle generated by the inverter(4) is three or smaller. It is therefore possible to ensure the accuracyof torque control (that is, to minimize an error between the torquecommand and the actual torque).

Also, the vector control device of an induction motor of the inventionis applied to a motor control device of an electric vehicle.Accordingly, it is possible to obtain a vector control system capable ofdriving an electric vehicle in a stable manner over a range from a lowspeed to a high speed where the output voltage of the invertersaturates. Also, it is possible to obtain a vector control device of aninduction motor capable of minimizing a loss of the inverter and makingthe inverter smaller and lighter and therefore suitable to an electricvehicle.

Also, a vector control method of an induction motor of the invention isa vector control method of controlling driving of an induction motor (6)via an inverter (4), including: computing a secondary magnetic fluxcommand to the induction motor (6) by taking a maximum voltage that theinverter (4) can generate into account on a basis of a torque commandfrom an external, a DC voltage to be inputted into the inverter (4), andan inverter angular frequency, which is an angular frequency of an ACvoltage to be outputted from the inverter (4); generating a q-axiscurrent command and a d-axis current command on a d-q axes rotatingcoordinate system in reference to a secondary magnetic flux of theinduction motor (6) on a basis of the torque command and the secondarymagnetic flux command; computing an output voltage that the inverter (4)is to output on a basis of the q-axis current command, the d-axiscurrent command, and a circuit constant of the induction motor; andcontrolling the inverter (4) for the inverter (4) to output the outputvoltage.

Accordingly, the secondary magnetic flux command is generated in a feedforward manner independently of the output voltage saturation state ofthe inverter. It is thus possible to provide a control method capable ofperforming stable vector control over the entire range from a low speedrange to a high speed range of the induction motor without using afeedback loop for generating the secondary magnetic flux command.

Also, a drive control device of an induction motor of the inventionincludes: an inverter (4) configured to control driving of an inductionmotor (6); secondary magnetic flux command computing means (40) forcomputing a secondary magnetic flux command to the induction motor (6)by taking a maximum voltage that the inverter (4) can generate intoaccount on a basis of a torque command from an external, a DC voltage tobe inputted into the inverter (4), and an inverter angular frequency,which is an angular frequency of an AC voltage to be outputted from theinverter (4); q-axis/d-axis current command generating means (8 and 9)for generating a q-axis current command and a d-axis current command ona d-q axes rotating coordinate system in reference to a secondarymagnetic flux of the induction motor (6) on a basis of the torquecommand and the secondary magnetic flux command; output voltagecomputing means (voltage non-interference computation portion 14) forcomputing an output voltage that the inverter (4) is to output on abasis of the q-axis current command, the d-axis current command, and acircuit constant of the induction motor (6); and voltage command/PWMsignal generating means (50) for controlling the inverter (4) for theinverter (4) to output the output voltage.

It is thus possible to obtain a drive control device capable ofcontrolling the driving of the induction motor in a stable manner overthe entire range from a low speed range to a high speed range withoutusing a feedback loop for generating the secondary magnetic fluxcommand.

INDUSTRIAL APPLICABILITY

The invention is useful in achieving a vector control device of aninduction motor capable of performing stable vector control over theentire range from a low speed range to a high speed range of aninduction motor without using a feedback loop for generating a secondarymagnetic flux command.

1. A vector control device that controls driving of an induction motorvia an inverter, comprising: a secondary magnetic flux command computingmodule that computes a secondary magnetic flux command for the inductionmotor by taking a maximum voltage that the inverter can generate intoaccount on a basis of a torque command from an external source, a DCvoltage to be input into the inverter, and an inverter angularfrequency, which is an angular frequency of an AC voltage to be outputfrom the inverter; a q-axis/d-axis current command generator thatgenerates a q-axis current command and a d-axis current command on a d-qaxes rotating coordinate system in reference to a secondary magneticflux of the induction motor on a basis of the torque command and thesecondary magnetic flux command; an output voltage computing module thatcomputes an output voltage that the inverter is to output on a basis ofthe q-axis current command, the d-axis current command, and a circuitconstant of the induction motor; and a voltage command/PWM signalgenerator for controlling the inverter to output the output voltage;wherein the secondary magnetic flux command computing module has: anoutput voltage maximum value computation portion configured to computethe maximum voltage that the inverter can generate on a basis of the DCvoltage applied to the inverter; a maximum voltage secondary magneticflux command computation portion configured to compute a maximum voltagesecondary magnetic flux command, which is a secondary magnetic fluxcommand to bring the maximum voltage that the inverter can generate andthe output voltage into coincidence in magnitude; and a lower-orderpreference portion configured to choose and output either the maximumvoltage secondary magnetic flux command or a pre-set rated secondarymagnetic flux command, whichever is the smaller, as the secondarymagnetic flux command.
 2. The vector control device of an inductionmotor according to claim 1, wherein: the maximum voltage secondarymagnetic flux command computation portion computes the maximum voltagesecondary magnetic flux command on a basis of the torque command and theinverter angular frequency.
 3. The vector control device of an inductionmotor according to claim 1, wherein: the maximum voltage secondarymagnetic flux command is computed in accordance with the followingequation${\Phi\; 2\; H^{*}} = \sqrt{\frac{{- A} + \sqrt{A^{2} - B}}{C}}$ where:A = 2 ⋅ R 1 ⋅ ω ⋅ Tm^(*) − VMmax²$B = {{4 \cdot \frac{\left\{ {{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}} \right\} \cdot \left\{ {{R\; 1^{2}} + {\sigma^{2}\left( {{\omega \cdot L}\; 1} \right)}^{2}} \right\}}{M^{4}} \cdot {Tm}^{*^{2}} \cdot L}\; 2^{2}}$$C = {2 \cdot \frac{{R\; 1^{2}} + \left( {{\omega \cdot L}\; 1} \right)^{2}}{M^{2}}}$and where VMmax is a maximum value of the output voltage of theinverter, Tm* is the torque command, ω is the inverter angularfrequency, R1 is a primary resistance of the motor, M is a mutualinductance of the motor, σ is a leakage coefficient, L1 is a primaryself-inductance of the motor, and L2 is a secondary self-inductance ofthe motor.
 4. The vector control device of an induction motor accordingto claim 1, wherein: the rated secondary magnetic flux command has atleast two kinds of values including a value applied during powering ofthe induction motor and a value applied during regeneration and isconfigured to be capable of switching the values according to a runningstate of the induction motor.
 5. The vector control device of aninduction motor according to claim 3, wherein: the rated secondarymagnetic flux command is a value set through preliminary computationusing the equation.
 6. The vector control device of an induction motoraccording to claim 1, wherein: a pulse mode of the inverter is switchedin response to percent modulation of the inverter computed on a basis ofthe secondary magnetic flux command and the torque command.
 7. Thevector control device of an induction motor according to claim 1,further comprising: a current detector configured to measure a currentflowing through the induction motor; a three-phase/d-q axes coordinatetransformer configured to convert the current detected by the currentdetector to a q-axis current and a d-axis current, which are values onthe d-q axes rotating coordinate system; a q-axis current control modulethat operates to lessen a deviation between the q-axis current commandand the q-axis current; and a d-axis current control module thatoperates to lessen a deviation between the d-axis current command andthe d-axis current, wherein: the output voltage computing modulecomputes the output voltage using outputs of the q-axis current controland the d-axis current control module and computations by the q-axiscurrent control module and the d-axis current control module are stoppedin a case where a half cycle generated by the inverter contains three orfewer pulses.
 8. The vector control device of an induction motoraccording to claim 7, wherein: the inverter angular frequency iscorrected using a deviation between the q-axis current command and theq-axis current in a case where the number of pulses in the half cyclegenerated by the inverter is three or less.
 9. The vector control deviceof an induction motor according to claim 1, wherein: the inverter isoperated in a single-pulse mode when percent modulation of the invertercomputed on a basis of the secondary magnetic flux command is 0.95 orhigher.
 10. The vector control device of an induction motor according toclaim 1, wherein: the vector control device is applied to a motorcontrol device of an electric vehicle.
 11. A vector control method ofcontrolling driving of an induction motor via an inverter, comprising:computing a secondary magnetic flux command for the induction motor bytaking a maximum voltage that the inverter can generate into account ona basis of a torque command from an external source, a DC voltage to beinput into the inverter, and an inverter angular frequency, which is anangular frequency of an AC voltage to be output from the inverter;generating a q-axis current command and a d-axis current command on ad-q axes rotating coordinate system in reference to a secondary magneticflux of the induction motor on a basis of the torque command and thesecondary magnetic flux command; computing an output voltage that theinverter is to output on a basis of the q-axis current command, thed-axis current command, and a circuit constant of the induction motor;and controlling the inverter to output the output voltage, wherein thestep of computing of the secondary magnetic flux command includes:computing the maximum voltage that the inverter can generate on a basisof the DC voltage applied to the inverter; computing a maximum voltagesecondary magnetic flux command, which is a secondary magnetic fluxcommand to bring the maximum voltage and the output voltage intocoincidence in magnitude; and choosing and outputting either the maximumvoltage secondary magnetic flux command or a pre-set rated secondarymagnetic flux command, whichever is the smaller as the secondarymagnetic flux command.
 12. A drive control device of an induction motor,comprising: an inverter configured to control driving of an inductionmotor; a secondary magnetic flux command computing module that computesa secondary magnetic flux command for the induction motor by taking amaximum voltage that the inverter can generate into account on a basisof a torque command from an external source, a DC voltage to be inputinto the inverter, and an inverter angular frequency, which is anangular frequency of an AC voltage to be output from the inverter; aq-axis/d-axis current command generator that generates q-axis currentcommand and a d-axis current command on a d-q axes rotating coordinatesystem in reference to a secondary magnetic flux of the induction motoron a basis of the torque command and the secondary magnetic fluxcommand; an output voltage computing module that computes an outputvoltage that the inverter is to output on a basis of the q-axis currentcommand, the d-axis current command, and a circuit constant of theinduction motor; and a voltage command/PWM signal generator forcontrolling the inverter to output the output voltage, wherein thesecondary magnetic flux command computing module has: an output voltagemaximum value computation portion configured to compute the maximumvoltage that the inverter can generate on a basis of the DC voltageapplied to the inverter; a maximum voltage secondary magnetic fluxcommand computation portion configured to compute a maximum voltagesecondary magnetic flux command, which is a secondary magnetic fluxcommand to bring the maximum voltage and the output voltage intocoincidence in magnitude; and a lower-order preference portionconfigured to choose and output either the maximum voltage secondarymagnetic flux command or a pre-set rated secondary magnetic fluxcommand, whichever is the smaller, as the secondary magnetic fluxcommand.